The demand of pulse radars and short-range sensors capable of detecting obstacles or levels of liquid with a resolution smaller than 0.1 m is increasing and manufacturers offer integrated radio frequency transceivers that operate in the microwave range. In a pulse radar system, short radio frequency pulses are transmitted to the antenna, are reflected by the obstacle and are then detected by the receiver. From the delay of the received echo, the distance of the target may be calculated by multiplying the speed of the light by half of the delay time.
The minimum detectable delay time equals the duration of the radio frequency pulse. As a consequence, the duration of the radio frequency pulse determines the minimum distance that the sensor may detect. Targets closer than this minimum distance would cause a superposition between the transmitted signal and the received echo signal.
Many techniques for generating very fast radio frequency pulses are available in literature. For instance, there are pulse radar systems that generate radio frequency bursts by switching on/off a high frequency oscillator that is generally realized with GaAs devices or Gunn oscillators as disclosed in U.S. Pat. Nos. 6,414,627 and 4,470,049. The oscillator is enabled for detecting the echo signal at the receiver side by controlling the input of the local oscillator either of the sampler or of the mixer.
The minimum detectable distance is determined by the switching time of the oscillator that is proportional to the attenuation time constant of the oscillator, and to the quality factor of the resonant circuit used for tuning the oscillator.
Generally, integrated radio frequency transceivers include a single reference low frequency oscillator and a PLL (Phase Locked Loop) for generating the high frequency pulse signal. In practice, the high frequency oscillator works in a continuous functioning mode and not in a discontinuous functioning mode, and the radio frequency bursts may be generated by introducing switches along the transmission path. This method has been disclosed in U.S. Pat. No. 5,239,309 by combining different spectral components of the pulse signal coming from different PLLs.
In transmissions with TDMA protocols that are largely used in wireless systems, switches are required. The so-called single-pole double-throw switches (SPDT) are largely used at the end of the transmission path. They are realized with diodes or with GaAs active devices or with CMOS devices. CMOS devices are preferred because they may not require input bias currents and have smaller costs and power consumption than the former ones.
A typical SPDT switch is shown in FIG. 1. It is controlled by the signal VCTRL that alternately connects the antenna to the TX transmission line or to the reception line RX. This switch has a low insertion loss of the signal to be transmitted when the respective switch is in a conduction state, and isolates the switch from the transmission line when the transistor is turned off. In the latter situation, the transmission line is connected to a reference potential and a strong impedance mismatch may be present.
Typically, a VCO coupled with a tuned reactive load generates an oscillating signal at a certain frequency on the transmission line. By switching the transmission line from the antenna and the reference potential, the line connected to ground functions as a reactive load in parallel to the tuned load that modifies the oscillation frequency of the VCO. Moreover, this load variation starts a transient that must be left to decay. Therefore, the pulses sent to the antenna should last at least a certain or minimum time. In addition, the selector switch of FIG. 1 needs a differential control signal and balancing this signal at high switching frequencies is not easy and typically requires a complex circuit.
A selector switch SPST with shorter switching times, disclosed in P. R. Gray and R. G. Meyer, Analysis and Design of Integrated Circuits, 2nd ed. New York: Wiley 1984, is depicted in FIG. 2. It is inserted at the beginning of the transmission line to the antenna, symbolically represented with a differential load LOAD. The depicted selector switch transfers the differential signal VIN+, VIN− to be transmitted to the load LOAD or to a dummy line, depending on the level of the differential control signal VCTRL+, VCTRL−.
This selector switch is burdened by drawbacks due to the Miller effect that limits the speed of the response of the transistors M2 and M3 and, as a consequence, the speed of the response to the variations of the input impedance between the on state and the off state. A remedy may include installing on the auxiliary line DUMMY LOAD an impedance (not depicted in the figure) equal to the load LOAD.
A generally known architecture of a selector switch that prevents variations of the input impedance seen by the local oscillator that generates the signal VIN is depicted in FIG. 3. It substantially exploits a so-called current steering technique for switching the single-ended input signal VIN on the load LOAD.
The performance of this type of selector switch is not limited by the Miller effect, because the transistors controlled by the differential control signal always see the same load whichever is the single-ended input voltage VIN.
Even in this case, a high frequency differential control signal VCTRL+, VCTRL− is necessary that is difficult to generate in a perfectly balanced manner. Moreover, switching times are determined exclusively by the pass band of the transistors of the differential pair.